A Methodology for multi-band class E RF PA design.

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A Methodology for multi-band class E RF PA design.

Antoine Diet, Martine Villegas, Geneviève Baudoin, Fabien Robert

To cite this version:

Antoine Diet, Martine Villegas, Geneviève Baudoin, Fabien Robert. A Methodology for multi-band

class E RF PA design.. IEEE International Microwaves Workshops Series on “RF front ends for

Software Defined and Cognitive Radio Solutions” IMWS 2010., Feb 2010, Aveiro, Portugal. p. n°7.

�hal-00553308�

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A Methodology for multi-band class E RF PA design

A. Diet

¹

, M. Villegas

²

, G. Baudoin

²

, F. Robert

²

1

UMR 8506, L2S-DRÉ. Supélec, F-91192 Gif Sur Yvette, France. antoine.diet@lss.supelec.fr

2

Université Paris-Est, ESYCOM, EA 2552; Groupe ESIEE, Noisy-Le-Grand 93162, France.

Abstract — This paper presents a simple methodology for the design of a highly efficient PA in the context of multi- radio devices. This PA belongs to the radiofrequency reconfigurable part within a transmitter. Our goal is to improve the PA efficiency over the bandwidth. The method is based on the extraction of the transistor’s non-linear output parameters and the optimisation of a given class E topology to an ideal impedance load. Herein, the method is illustrated by an integrated design using between 2 and 4 GHz and the results of the simulations (in terms of efficiency) agree well with the transient simulation and the S-Parameters optimisation. Considerations are given for the complete development of this design and the interest for future multi- band front end designs.

Index Terms — RF architecture, multi-radio, class E PA.

I. THE MULTI-RADIO CONTEXT

Wireless communications are widely used for our daily needs. There are numerous examples which use between 2 and 4 GHz and standard names like BLUETOOTH, WiFi, WiMAX, UMTS and LTE are well-known. An important aspect of today’s applications is the increasing data rate need, especially in connectivity standards (WiFi, WiMAX), because of the user’s high Quality of Service (QoS) demands. To increase the data rate, we tend to use wideband or multi-standard architecture [1] [5]. The concept of software radio includes a self-reconfigurable radio link. While focusing on the radiofrequency (RF) part, the term multi-radio is preferred. A multi-radio RF device is supposed to use different RF flexible blocks, whose parameters are the centre/carrier frequency, bandwidth, modulation scheme and average power, see figure 1.

figure 1 : Multi-radio front-end challenges

Also, the signal dynamics (represented by the Peak to Average Power Ratio) should not decrease the transmitter’s performance. Due to the different PAPRs of the standard signals (OFDM for example), the transmitter has to be linearised, because the power amplifier (PA) introduces crippling non-linear effects (NL) [13] [4]. The

present challenge is to linearise the RF architecture while keeping high efficiency, because this is related to battery- life (especially in our context of nomadic transceiver design). Lots of linearization techniques were proposed [1] and recent works made EER/polar-based architecture very popular [6]. The envelope information can be coded by a Pulse Width Modulation or a Sigma Delta (ΣΔ) process in order to present a constant power property which is beneficial for the architecture efficiency. This coding enables its recombination by a multiplication with the carrier signal, which is modulated by the phase information. This multiplication can be achieved before the PA or by supply modulation, if the PA is in a switched (SW) class [4] such as class D, S, E or F [3].

As RF blocks are flexible, the size of the transmitter is optimized and the system efficiency is directly dependant on the different configurations. Moreover, a major design challenge is to keep a highly efficient PA for the different bands specified by the standards (carrier frequency and bandwidth). The idea of the front end design is an optimization (frequency sweeping) of a Class E PA. The Class E was chosen for its high efficiency performance and for the small number of reactive components needed, whatever the topology is [3] [7] [8] [9] [10] [11] [12] [13].

As was discussed in [2], PAs for multi-radio can be of three types: (i) broadband, (ii) reconfigurable (for example: [5]) or (iii) multi-band matching PAs. We present in this paper a simple methodology based on previous studies of Class E and optimisation processing. A slight increase in the number of reactive components can improve the bandwidth of such a PA and drive us to a dual band PA optimisation.

II. PROPOSED DESIGN METHOD

The proposed design method is summarized in figure 2.

Once the given transistor and frequency bands are chosen, the idea is to first extract the non-linear (NL) output of the transistor in switched mode of operation, in part one of the method. In that order, we performed a transient simulation where the transistor input voltage implies a hard switching behaviour (a FET is supposed here). This hypothesis is achieved by a closed loop on the gate-to- source voltage that maintains the saturation of device whatever the frequency is, see figure 2. Thanks to this simulation, we calculated the frequency-dependant gate-

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to-source voltage, and the resulting input power Pin (un- matched case). A Large Signal S-Parameters (LSSP) frequency simulation, including the parametric input power Pin, computes the LSSP output matching coefficient (S22 NL) of the transistor. In part two of the method, we model the real transistor as an ideal switch with its S22 NL (impedance) in parallel: a transistor and a

capacitor, as is drawn in figure 2 and confirmed by the Smith chart. This supposes that we consider the unilaterality (S12 = 0).

At this point, we can compute the load facing the ideal switch, made up of the S22 NL, the output network (to be designed) and the antenna, as seen in Figure 2.

OPTIMISATION, with the definition of Z_opt

Simulation of the transistor in switched mode over the bandwidth and computation of the corresponding input power (transient simulation)

Characteristics Vgs, Ids, Vds of the transistor Definition of the gate voltage for switched mode at each frequency

Pin

Simulation in LSSP (Pin) Large Signal S-Parameter (frequency simulation)

S22

Extraction of S22, and definition of a LC topology, loaded by the antenna.

(S-Parameter frequency simulation)

Simulation of the optimum LC network with the transistor in switched mode (transient simulation). Computation of the efficiency.

Feedback loop

Input Power

VgsFREQ (2000000000.000 to 4000000000.000)

S(2,2)

(0.000 to 0.000)

G_OPT

freq (2.000GHz to 4.000GHz)

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

S22 = Zout Zload S22 + Zload

EqnZ_OPT=(31.155+j*23.265) (0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

V_DC I_drain

Z_load L_pola Output

Network

RF08265 load ^j³⁴^

opt f . R e

Z fRF

Real transistor Output Network with DC feed Antenna

Ideal transistor

NL output impedance

V_drain OPTIMISATION, with

the definition of Z_opt

Pin

S22

OPTIMISATION, with the definition of Z_opt

Pin

S22

Feedback loop Feedback loop

Input Power

VgsFREQ (2000000000.000 to 4000000000.000)

S(2,2)

(0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

VgsFREQ (2000000000.000 to 4000000000.000)

S(2,2)

(0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

S22 = Zout Zload S22 + Zload

EqnZ_OPT=(31.155+j*23.265) (0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

EqnZ_OPT=(31.155+j*23.265) (0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

(0.000 to 0.000)

G_OPT

S(1,1)

V_DC I_drain

Network

RF08265 load ^j³⁴^

opt f . R e

Z fRF

V_drain

V_DC I_drain

Network

RF08265 load ^j³⁴^

opt f . R e

Z fRF

V_drain

Figure 2: Proposed design method Based on numerous theoretical developments

concerning class E topologies, [13] [12] [7] [8] [9] [10]

and [11], we focused on parallel-based topologies with finite DC feed inductor. The ideal load that must face the ideal transistor (the “perfect switch” part) of this type of topology [5] [11] is Zopt(ωRF) = 0.8265 Rload e^j34°, where Rload is the reactive part of the antenna (radiating part) and ωRF the centre/carrier frequency. The optimisation process goals are to match the impedance of the circuit

(S22 NL + output network + antenna) to the optimal class E load Zopt. We compute the matching coefficient ГclassE according to this criteria (50 Ohms is replaced by Z0

= Zopt = 0.8265 Rload e ^j34°). Once the optimisation process meets the goals, the network is incorporated into the final transient simulation in order to evaluate the resulting efficiency over the bandwidth.

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We applied this method with the model of a MESFET Avago ATF50189 and a designed antenna (Vivaldi type).

Results of the S22 NL (Zout) extraction are reported in Figure 2 (left) and the antenna impedance is drawn in the left, middle smith chart. Three networks are computed: for 2 to 2.5 GHz (LTE, UMTS band), 3.2 to 3.8 GHz (WiMAX) and for both frequency bands. Figure 3 shows the results of the optimisation for the three networks, by computing the matching coefficient ГclassE . We reported the result of the final transient simulation only with the network matching both frequency bands. This networks uses one DC feed inductor of 1 nH, two grounded shunt inductors of 1.62 and 2.36 nH, two series capacitors of 0.8 and 0.9 pF and two shunt capacitors of 0.4 and 1 pF. The

efficiency and output power are drawn in Figure 3. We notice high theoretical drain efficiency in the range of 90%, corresponding to the goal parameters of our optimisation process. In practice, this will be lowered by the practical loss of the circuit. Maximum efficiencies agree well with matched frequency bands predicted by the S-Parameters simulation.

Drain to source voltages and currents are drawn for 2.3 and 3.5 GHz, in order to demonstrate the class E operation of the amplifier. We notice the negative current due to the importance of the output transistor capacitance (intrinsic) at these frequencies of operation.

2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

2.0 4.0

-25 -20 -15 -10 -5

-30 0

freq, GHz

dB(G_ADAPT)dB(S(1,1))

2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9

2.0E9 4.0E9

13 16 19 22 25 28

10 30

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9

0.0 1.0

VgsFREQ

10*log10(P*1000) ETA

2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

2.0 4.0

-25 -20 -15 -10 -5

-30 0

freq, GHz

2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9

2.0E9 4.0E9

13 16 19 22 25 28

10 30

0.6 0.7 0.8 0.9

0.5 1.0

VgsFREQ

10*log10(P*1000) ETA

2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9

2.0E9 4.0E9

13 16 19 22 25 28

10 30

0.6 0.7 0.8 0.9

0.5 1.0

VgsFREQ

10*log10(P*1000) ETA

2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

2.0 4.0

-40 -30 -20 -10

-50 0

freq, GHz

-2 0 2 4

-4 6

freq_optim2bis..Vcharge3, V

-50 0 50

-100 100

freq_optim2bis..Icharge.i, mA

0 2 4 6 8

-2 10

freq_optim2bis..Vdrain3, V

1. 000 1. 003

-0. 10 -0. 05 0. 00 0. 05 0. 10 0. 15 0. 20

-0. 15 0. 25

time, us ec

-freq_optim2bis..I_Probe3.i

2.3 GHz

-4 -2 0 2 4

-6 6

-50 0 50 100

-100 150

0 2 4 6 8

-2 10

800 801

-0.2 -0.1 0.0 0.1 0.2 0.3

-0.3 0.4

time, nsec

3.5 GHz

2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

2.0 4.0

-25 -20 -15 -10 -5

-30 0

freq, GHz

2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9

2.0E9 4.0E9

13 16 19 22 25 28

10 30

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9

0.0 1.0

VgsFREQ

10*log10(P*1000) ETA

2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

2.0 4.0

-25 -20 -15 -10 -5

-30 0

freq, GHz

2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9

2.0E9 4.0E9

13 16 19 22 25 28

10 30

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9

0.0 1.0

VgsFREQ

10*log10(P*1000) ETA

2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

2.0 4.0

-25 -20 -15 -10 -5

-30 0

freq, GHz

2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9

2.0E9 4.0E9

13 16 19 22 25 28

10 30

0.6 0.7 0.8 0.9

0.5 1.0

VgsFREQ

10*log10(P*1000) ETA

2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

2.0 4.0

-25 -20 -15 -10 -5

-30 0

freq, GHz

2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9

2.0E9 4.0E9

13 16 19 22 25 28

10 30

0.6 0.7 0.8 0.9

0.5 1.0

VgsFREQ

10*log10(P*1000) ETA

2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9

2.0E9 4.0E9

13 16 19 22 25 28

10 30

0.6 0.7 0.8 0.9

0.5 1.0

VgsFREQ

10*log10(P*1000) ETA

2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

2.0 4.0

-40 -30 -20 -10

-50 0

freq, GHz

2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9

2.0E9 4.0E9

13 16 19 22 25 28

10 30

0.6 0.7 0.8 0.9

0.5 1.0

VgsFREQ

10*log10(P*1000) ETA

2.2 2.4 2.6 2.8 3.0 3.2 3.4 3.6 3.8

2.0 4.0

-40 -30 -20 -10

-50 0

freq, GHz

-2 0 2 4

-4 6

-50 0 50

-100 100

0 2 4 6 8

-2 10

1. 000 1. 003

-0. 10 -0. 05 0. 00 0. 05 0. 10 0. 15 0. 20

-0. 15 0. 25

time, us ec

2.3 GHz

-4 -2 0 2 4

-6 6

-50 0 50 100

-100 150

0 2 4 6 8

-2 10

800 801

-0.2 -0.1 0.0 0.1 0.2 0.3

-0.3 0.4

time, nsec

3.5 GHz

-2 0 2 4

-4 6

-50 0 50

-100 100

0 2 4 6 8

-2 10

1. 000 1. 003

-0. 10 -0. 05 0. 00 0. 05 0. 10 0. 15 0. 20

-0. 15 0. 25

time, us ec

2.3 GHz

-4 -2 0 2 4

-6 6

-50 0 50 100

-100 150

0 2 4 6 8

-2 10

800 801

-0.2 -0.1 0.0 0.1 0.2 0.3

-0.3 0.4

time, nsec

3.5 GHz

Figure 3 : Switched mode simulation of the transistor (transient) for three types of NT : (i) single band WiFi, (ii) single band WiMAX a nd (iii) dual band WiFi/WiMAX with transient drain to source voltage and current at 2.3 and 3.5 GHz. Efficiency (ETA in purple) and Output power (red) are reported on the same figure. Matching coefficient for 50 Ohms (blue) and Zopt (red) are draw with frequency alignment.

The drain-to-source voltage is varied from 1 to 3 V and shows the independence of the efficiency and the quasi- linear variation of the output power, characterizing the saturation of the transistor. These results are shown in figure 4. It is important to evaluate this efficiency of the dual band PA over the voltage and the frequency because

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the multi-radio idea implies a power control whatever the signal parameters are.

  





  ₃₄

34

826 0

j load

-j load with R

class E

e R . Z

e R . Z Γ Z

load 2.0E9 2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9 4.0E9

13 16 19 22 25 28

10 30

0.6 0.7 0.8 0.9

0.5 1.0

VgsFREQ

10*log10(P*1000) ETA ETA1 ETA2

10*log10(P1*1000)10*log10(P2*1000)

(0.000 to 0.000)

G_OPT

S(1,1)

Г_classEin dB

S₁₁in dB

Poutin dBm Efficiency(normalised)Vds = 1 V

Vds = 3 V

Vds = 2 V

  





  ₃₄

34

826 0

j load

-j load with R

class E

e R . Z

e R . Z Γ Z

load 2.0E9 2.2E9 2.4E9 2.6E9 2.8E9 3.0E9 3.2E9 3.4E9 3.6E9 3.8E9 4.0E9

13 16 19 22 25 28

10 30

0.6 0.7 0.8 0.9

0.5 1.0

VgsFREQ

10*log10(P*1000) ETA ETA1 ETA2

10*log10(P1*1000)10*log10(P2*1000)

(0.000 to 0.000)

G_OPT

S(1,1)

Г_classEin dB

S₁₁in dB

Poutin dBm Efficiency(normalised)Vds = 1 V

Vds = 3 V

Vds = 2 V

Figure 4 : Results of transient simulation with the transistor and the optimized dual-band network: efficiency and output power in

function of the frequency and for Vds = 1, 2 and 3 V

III. CONCLUSION

We presented a simple methodology for class E efficiency optimisation over frequency. This method helps drive the study of broadband or multi-band PA for multi- radio. Results based on a real transistor model reach almost a one octave class E PA with reactive element values in an acceptable range for integrated design. The small number of reactive component is attractive for the design of a dual-band PA, as presented. The possibility of output power variations are simulated for slow (average power) or fast (envelope information) amplitude control.

We will extend the number of frequency bands according to multi-radio front end needs.

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